The present invention relates to a switching power supply device which supplies prescribed output power to a load according to a preset voltage, and in particular relates to a switching power supply device which lowers the switching frequency of a switching element during light loading or no loading or standby (hereinafter also simply called “light loading”), thereby achieving reduced power consumption during light loading or no loading, or reduced standby power during standby.
In the prior art, IC circuits for switching power supply control have been utilized which improve power efficiency by lowering the switching frequency during-light loading, with the aim of lowering switching losses in the switching power supply (see for example Japanese Patent Laid-open No. 2007-215316 (page 1, line 15 to page 13, line 16, FIG. 4, FIG. 5)). In Japanese Patent Laid-open No. 2007-215316 (page 1, line 15 to page 13, line 16, FIG. 4, FIG. 5), a switching power supply control circuit is disclosed employing current mode control (positive detection method), in which the value of the current flowing in the power MOSFET (Metal-Oxide Semiconductor Field Effect Transistor) or other switching element is detected as a positive voltage signal.
FIG. 9 is a block diagram showing the control circuit of a quasi-resonant switching power supply disclosed in Japanese Patent Laid-open No. 2007-215316 (page 1, line 15 to page 13, line 16, FIG. 4, FIG. 5). The quasi-resonant switching power supply of FIG. 9 is shown as only one example of a switching power supply of the prior art. This invention is not limited to quasi-resonant devices, and can also be applied broadly to switching power supplies other than quasi-resonant devices.
In the power supply control circuit 10, a bottom detection circuit (valley detection) 11 is connected to the input terminal ZCD for zero current detection. The bottom detection circuit 11 is a comparator which compares the voltage applied to the input terminal ZCD with a reference voltage close to the 0 V level (threshold); the output terminal of this bottom detection circuit 11 is connected to one of the input terminals of the AND circuit 12, and through the AND circuit 12, is connected to the one-shot circuit 13. The output terminal of the voltage controlled oscillator (VCO) 14 is connected to the other input terminal of the AND circuit 12. The voltage controlled oscillator 14 is an oscillator which changes the output frequency depending on the magnitude of the input voltage (VCO voltage), and comprises a voltage signal input terminal VCO and a reset signal input terminal Reset. The VCO voltage input terminal of the voltage controlled oscillator 14 is connected to the input terminal FB for feedback signal VFB detection, and the input terminal for reset signals Reset is connected to the output terminal of the one-shot circuit 13.
The input terminal FB for detection of the feedback signal VFB is connected to the inverting input terminal (−) of the comparator 15. The non-inverting input terminal (+) of the comparator 15 is grounded via 0.5 V reference power supply E1, and a disable signal Disable is output from the output terminal to the inverter circuit 16. The output of the inverter circuit 16 is connected to the clear terminal (CLR) of the one-shot circuit 13. A 5 V reference power supply E2 is connected via the series circuit of a resistor R and diode D to the input terminal FB, and determines the FB terminal voltage.
The current comparator 17 is connected to the signal input terminal for current detection IS, and a current detection signal is supplied to the non-inverting input terminal (+) among the four input terminals of the current comparator 17. The remaining three inverting input terminals (−) are connected respectively to the input terminal for feedback signal VFB detection FB, the 1 V reference power supply E3, and the output terminal of the soft-start circuit 18. The output terminal of the current comparator 17 is connected to the reset terminal R of the flip-flop circuit 19. The set terminal S of the flip-flop circuit 19 is connected to the output terminal of the one-shot circuit 13. The Q output terminal of the flip-flop circuit 19 is connected via the AND circuit 20 to the output terminal OUT, and the output signal Q of the flip-flop circuit 19 is output to the power MOSFET or other externally connected switching element Q1 (see FIG. 10 below) as a switching signal from the output terminal OUT. The soft-start circuit 18 generates a soft-start signal, which limits the turn-on interval of the switching element Q1 during startup of the switching power supply.
The inverting input terminal (−) of the comparator for overload detection 21 is connected to the input terminal FB for feedback signal VFB detection, and the non-inverting input terminal (+) is grounded via the 3.3 V reference power supply E4. The output terminal of the comparator 21 is connected to the reset terminal Reset of the timer circuit 22. The timer circuit 22 is used to set two delay times; the first output signal (low) is output to the AND circuit 20 100 ms after the comparator 21 detects an overload state, and forcibly halts the supply of the switching signal to the switching element Q1.
The second output signal of the timer circuit 22 is output 800 ms after an overload state is detected, and is supplied as a reset signal to a startup circuit, not shown, provided within the power supply control circuit 10.
In this power supply control circuit 10 for a switching power supply disclosed in Japanese Patent Laid-open No. 2007-215316 (paragraphs [0002] to [0025], FIG. 4, FIG. 5), the voltage applied to the switching element Q1 upon zero-cross detection is the minimum of the resonance waveform, and the switching element Q1 is turned on with this timing to start the next switching cycle, in what is generally called a quasi-resonant type or partial-resonance type switching power supply control.
In the control circuit shown in FIG. 9, during normal operation the current signal of the switching element Q1 is input to the input terminal IS, and the current comparator 17 compares this current signal with the feedback signal VFB input to the input terminal FB; the power supplied to the secondary side is controlled by making the current in the switching element Q1 small when the load is light and making the current in the switching element Q1 large when the load is heavy, executing control so that the output voltage is substantially equal to the voltage setting.
The feedback signal VFB input to the input terminal FB decreases when the load is light and the output voltage is high, and increases when the load is heavy and the output voltage declines. The voltage controlled oscillator 14 lowers the frequency more for a smaller feedback signal VFB, which is the VCO voltage, so that the lighter the load, the lower the oscillation frequency of the voltage controlled oscillator 14, and the heavier the load, the higher the frequency. A detailed explanation is here omitted, but the frequency of the switching signal (the switching frequency) output from the output terminal OUT of the power supply control circuit 10 is governed by the oscillation frequency of the voltage controlled oscillator 14, so that, in essence, the lighter the load, the lower is the switching frequency. This is because in light loading, the fraction of total losses accounted for by switching loss is increased, and so the frequency is lowered with the aim of alleviating the switching loss during light loading. This technique of lowering the switching frequency during light loading is also widely applied to switching power supplies other than quasi-resonant type devices.
The reference voltage E3 (1 V) connected to the current comparator 17 is a reference voltage used to limit overcurrents in the switching element Q1. In the case of an overload and the like, the maximum value of the current signal is limited to the reference voltage E3 (1 V) in order to protect the power supply circuit and the load.
FIG. 10 is a block diagram showing one example of a positive detection-type switching power supply device of the prior art.
The switching power supply device of FIG. 10 supplies power from the primary-side DC input power supply VIN of a transformer T1 to the secondary-side load (not shown) according to a voltage setting. An LC resonance circuit is formed by the inductance (Lp) of the primary-side windings Lp of the transformer T1 and the capacitance of the resonance capacitor Cr (which can also be only the parasitic capacitance of the switching element Q1) connected in parallel with the power MOSFET or other switching element Q1. The input voltage VIN is supplied to one end of the smoothing capacitor C1 and to one end of the primary windings Lp of the transformer T1; the other end of the primary windings Lp is connected to the drain of the switching element Q1. The source of the switching element Q1 is connected, via the sense resistor Rs, to the other end of the smoothing capacitor C1, and the gate is connected via the resistance R1 to the output terminal OUT of the integrated circuit IC.
The integrated circuit IC in the switching power supply circuit of FIG. 10 is for example equivalent to the power supply control circuit 10 of FIG. 9; in FIG. 9, only the zero current detection input terminal ZCD, feedback signal detection input terminal FB, signal input terminal for current detection IS, and the output terminal OUT for output of the control signal to the switching element Q1 are shown.
The primary windings Lp, secondary windings Ls, and auxiliary windings Lb of the transformer T1 are all wound around the same core of the transformer T1. The inductance of the secondary windings Ls is Ls, and the inductance of the auxiliary windings Lb is Lb. The resonance capacitor Cr is connected in parallel with the series circuit of the switching element Q1 and the sense resistor Rs, but may be connected in parallel with the primary windings Lp. The auxiliary windings Lb are connected to a rectifying diode D2 and smoothing capacitor C2 which form the power supply of the integrated circuit IC. The resistor R2 supplies the voltage at the connection point between the switching element Q1 and the sense resistor Rs to the signal input terminal for current detection IS; and the resistor R3 is provided to input the voltage across the auxiliary windings Lb as-is, without rectification, to the input terminal ZCD of the integrated circuit IC. The sense resistor Rs functions as a current detection element.
A diode D3 and smoothing capacitor C3 which rectify the voltage appearing across the secondary windings Ls are provided at the secondary windings Ls of the transformer T1. The anode of the diode D3 is connected to one end of the secondary windings Ls, and the cathode is connected to the power supply output terminal Vout as well as to one end of the smoothing capacitor C3. The other end of the smoothing capacitor C3 is connected to the other end of the secondary windings Ls, as well as to the ground terminal Gnd.
The level at the output terminal OUT of the integrated circuit IC changes between high and low to drive the gate of the switching element Q1, turning the switching element Q1 on and off, and by this means the desired smoothed DC voltage is generated on the side of the secondary windings Ls of the transformer T1, between the power supply output terminal Vout and the ground terminal Gnd. During on intervals, a drain current flows in the switching element Q1, and a current flows on the side of the primary windings Lp of the transformer T1 connected thereto, accumulating energy. Thereafter the switching element Q1 turns off, but by means of the energy accumulated in the transformer T1, a current passes through the diode D3 and flows in the smoothing capacitor C3 on the side of the secondary windings Ls of the transformer T1 during off intervals of the switching element Q1. In this way, a smoothed DC voltage appears on the side of the secondary windings Ls of the transformer T1, between the power supply output terminal Vout and the ground terminal Gnd.
Between the power supply output terminal Vout and the ground terminal Gnd is provided an output detection circuit comprising a series circuit of resistors R5 and R6, a resistor R7, a photodiode PD comprising a phototransistor PT and a photocoupler, a capacitor C4, and a shunt regulator D4. Here, a current flows in the photodiode PD according to the output voltage (the current which flows is larger to the extent that the output voltage is higher than the voltage setting), the photodiode PD emits light in a quantity corresponding to this current, and a feedback signal is supplied to the phototransistor PT connected between the feedback signal detection input terminal FB of the integrated circuit IC and the ground terminal Gnd. The larger the quantity of light emitted by the photodiode PD, the larger the current which flows in the phototransistor PT, and this current flows in the resistor R, so that the voltage drop across the resistor R increases. That is, the higher the output voltage, the larger is the current flowing in the phototransistor PT, so that the feedback signal VFB becomes small. By means of this feedback function, a switching power supply device can supply power according to fluctuations in the load, not shown. The feedback circuit 25 comprises the portions enclosed within the dashed line.
The positive detection switching power supply device shown in FIG. 10 has a sense resistor Rs as a current detection element, and is characterized in comprising an overload protection (OLP; also called overcurrent protection, OCP) function, which protects the load from excessive currents by means of an overcurrent limiting circuit which employs the signal obtained by applying a bias, by means of the resistors R4 and R2, to the current detection signal (the signal itself being a voltage) detected by the sense resistor Rs. In recent devices, reduced power consumption in the power supply control circuit 10 itself has been sought, and a method of reducing the current flowing in the path from the input power supply VIN through the resistors R4, R2, Rs is conceivable. Before explaining this method, first the function of the resistors R4 and R2 is explained.
First, a state in which the resistors R4, R2 are not provided is considered. This overcurrent limiting circuit does not directly monitor overcurrent on the secondary side of the transformer T1, but instead monitors current changes on the side of the primary windings Lp to detect overcurrents to the load and halt switching operation. This is because when directly monitoring the secondary-side load current, a circuit to feed back a signal to the primary side becomes necessary. Specifically, the voltage across the sense resistor Rs which is the current detection element in FIG. 10 is compared with a reference voltage which is an overload protection (OLP) judgment reference (and is hereinafter called the judgment reference voltage Vth), and when the voltage across the sense resistor Rs reaches this reference voltage, it is judged that an overcurrent has occurred.
FIGS. 11(A) and 11(B) show the primary-side current waveforms corresponding to different input voltages. Here, when the respective input voltages VIN are applied as V1 and V2, the positive-detection current detection signal occurring across the resistor Rs is shown as the current waveform corresponding to the inductor current IL flowing in the primary windings Lp of the transformer T1.
An inductor current IL begins to flow in the primary windings Lp at the time that the switching element Q1, an N-channel MOS transistor, is turned on, and this current increases with a slope proportional to the input voltage VIN (dIL/dt=VIN/Lp). When the current detection signal reaches the judgment reference voltage Vth for overload protection (OLP), the power supply control circuit 10 (integrated circuit IC) of FIG. 9 judges that an overcurrent is occurring, and the switching element Q1 is turned off.
In the positive-detection type switching power supply device shown in FIG. 10, a response lag time Δt occurs between the time an overcurrent is actually judged to have occurred in the integrated circuit IC and the time the switching element Q1 is turned off. For this reason, as shown in FIGS. 11(A) and 11(B), overshoot exceeding the judgment reference occurs in the inductor current IL actually flowing in the switching element Q1 at the time of overcurrent limiting operation. While the slope of the inductor current IL is proportional to the input voltage VIN, the response lag time Δt, which is determined by operation of the control system, is regulated by the power supply voltage of the power supply control circuit 10 (the integrated circuit IC in FIG. 10), so that it is not affected by the input voltage VIN. Hence upon comparing the current detection signals from the sense resistor Rs for a case in which the input voltage VIN is a small value (V1) as shown in FIG. 11(A), and for a case in which the value is V2 (>V1) shown in FIG. 11(B), the above-described overshoot amount ΔV is larger for higher values of the input voltage VIN (ΔV1<ΔV2).
FIG. 12 shows changes in the inductor current during overcurrent limiting operation in the switching power supply device of FIG. 10. When the switching element Q1 is cut off after a lag time, that is, during overcurrent limiting operation, the inductor current IL flowing in the primary windings Lp of the transformer T1 increases in proportion to the input voltage VIN, as explained in FIGS. 11(A) and 11(B). In a conventional positive-detection type switching power supply device, when for example the device is used as the power supply for a personal computer in Japan, the 100 V commercial AC power supply is rectified and smoothed for use as the DC input power supply. In other countries, a 200 V AC power supply may be used. On the other hand, only voltages of at most approximately 10 to 20 V are required as the output voltages from the secondary windings Ls or from the auxiliary windings Lb of the transformer T1. When there is deviation in the voltage of the commercial AC power supply, that is, in the input voltage VIN, if the higher the input voltage VIN the larger the inductor current IL which flows when turning off the switching element Q1, problems are posed for power supply safety.
Hence with the aim of correcting for this overshooting in the switching power supply device shown in FIG. 10, a resistance circuit in which resistors R2 and R4 are connected in series is provided. By means of this resistance circuit, the voltage level of the sense resistor Rs is shifted in the positive direction. The amount of level shifting is larger for higher input voltages VIN, so that the higher the input voltage VIN, the more quickly an overcurrent state can be judged in the stage before the voltage of the sense resistor Rs reaches the overcurrent limiting judgment reference voltage Vth. Hence the overshoot amount ΔV when actually turning off the switching element Q1 can be compensated by this resistance circuit.
However, in a positive detection method in which the level is shifted by a resistance circuit, when considered from the standpoint of reduction of power consumption under light loading or no loading or reduction of standby power during standby, which has emerged as an issue in power supply systems in recent years, the power consumption due to the current flowing from the input power supply VIN (in a normal power supply system, the input power supply VIN is at the highest voltage) through the resistance circuit of the resistors R4, R2, Rs to ground (Gnd) becomes a problem. Hence a method is known in which, in order not to pass unnecessary current in a switching power supply device, while compensating for the phenomenon in which the higher the input voltage VIN, the larger is the overcurrent when the switching element Q1 is turned off, as shown in FIG. 12, the current detection signal level is shifted in the negative direction, to achieve reduced power consumption (see for example Japanese Patent Laid-open No. 2003-299351 (page 8, line 2 to page 9, line 24, FIG. 2)).
FIG. 13 is a block diagram showing one example of a negative-detection type switching power supply device of the prior art.
As for example in the case of the switching power supply device disclosed in Japanese Patent Laid-open No. 2003-299351 (page 8, line 2 to page 9, line 24, FIG. 2), a negative-detection type switching power supply device is configured such that current detection means employs a sense resistance Rs to detect the current flowing in the primary windings, or the current flowing in the switching element, as a negative voltage. Hence in the switching power supply device shown in FIG. 13, the signal input terminal for current detection IS and the sense resistor Rs are connected via a resistor Ra. Moreover, the signal input terminal IS is connected to the connection point between the auxiliary windings Lb and the rectifying diode D2, and to the power supply terminal VCC supplying power to the integrated circuit IC, via the resistor Rb and the correction resistor Rc, respectively.
First, the functions of the resistors Ra and Rb are explained (the function of the correction resistor Rc is explained below). As is clear from the circuit configuration in FIG. 13, the larger the current on the primary side, the larger the absolute value of the negative voltage which becomes the current detection signal. The resistors Ra and Rb correspond to the respective resistors R2 and R4 in the positive detection method of FIG. 10, and are provided to apply a negative bias to the current detection signal. During intervals in which the switching element Q1 is turned on, a negative potential appears at the connection point between the auxiliary windings Lb and the rectifying diode D2. This negative potential is insulated from the smoothing capacitor C2 by the rectifying diode D2, and so is proportional to the input voltage VIN (but with the sign inverted). Hence just as when the positive-voltage current detection signal has a positive bias applied in the positive detection method, so in the negative detection method the negative-voltage current detection signal has a negative bias applied, which is proportional to the input voltage VIN.
In the positive detection method and the negative detection method, the power consumed in the resistors R4, R2, Rs and in the resistors Rb, Ra, Rs differs greatly. This is because the power consumed in a resistor is proportional to the square of the voltage applied to the resistor ((voltage)2/resistance value), and the applied voltages differ greatly. As explained above, when a commercial AC power supply is rectified and smoothed to obtain an input voltage VIN, the value is approximately 100 to 200 V, whereas the output voltage (absolute value) from the auxiliary windings Lb is at most approximately 10 to 20 V, so that the power consumption can be reduced by about two orders of magnitude.
In the power supply control circuit 10 (IC circuit) of FIG. 13, only a portion of the elements comprised by the circuit is shown. Here, the voltage controlled oscillator 14, current comparator 17, and flip-flop circuit 19 are circuits corresponding to the control circuit shown in FIG. 9, a signal inversion circuit 23 is positioned to supply signals to the non-inverting input terminal (+) of the current comparator 17 from the feedback signal VFB detection input terminal FB, and a level shift circuit 24 is provided between the current detection input terminal IS and the inverting input terminal (−) of the current comparator 17. Although omitted in FIG. 13, the power supply control circuit 10 also comprises a zero current detection input terminal ZCD, a terminal VH to which startup current is supplied, and the like.
The voltage controlled oscillator 14 is an oscillator used to determine the switching frequency; and the oscillation frequency is controlled by a feedback signal VFB (this signal is equivalent to a so-called error signal), output from the feedback circuit 25, resulting from amplification of the difference between the voltage output to the load and the voltage setting. The frequency characteristic is such that, in the range in which the load is judged to be light (for example, when the feedback signal VFB is 0.9 V or less), the frequency is proportional to the voltage of the feedback signal VFB, and declines substantially linearly to the minimum frequency. When the load is heavy, the frequency is constant (the maximum frequency). The feedback circuit 25 is the same as that shown in FIG. 10.
The larger the feedback signal VFB, the heavier the load is judged to be, so that increasing the output current such that the output voltage reaches the target voltage setting is difficult, and so the switching frequency is raised so as to enable accommodation of large changes in the load current. And the smaller the feedback signal VFB, the lighter the load with a small output current is judged to be, so that the switching frequency is set low.
When the feedback signal VFB is smaller than a prescribed value (for example 0.4 V), switching is stopped, and a feedback signal VFB voltage higher than the above prescribed value of 0.4 V is awaited. No switching is performed, so that electric charge is not supplied to the secondary-side output capacitor C3, and current is supplied only to the load, so that the output voltage falls. As a result the difference between the output voltage and the voltage setting increases, and the voltage value of the feedback signal VFB rises.
FIG. 14 shows the configuration of the signal inversion circuit 23 of the switching power supply device shown in FIG. 13. The signal inversion circuit 23 comprises an operation amplifier circuit 26, resistors R11 and R12, and a reference voltage supply E5, as shown in FIG. 14.
Here, the feedback signal VFB is supplied from the feedback circuit 25 via the input terminal FB as a voltage signal of 1 to 2 V, suitable for the positive detection method. The signal is inverted and amplified by the signal inversion circuit 23, to be converted into an internal signal VFB2 of 2 to 1.5 V conforming to the negative detection method. The voltage values used by the signal inversion circuit 23 are examples used to explain the range of values which signals may take, and signals are not limited to these values.
FIG. 15 shows the configuration of the level shift circuit 24 in the switching power supply device shown in FIG. 13. The level shift circuit 24 comprises a resistor R13 for protection from static electricity and a series circuit of resistors R14, R15 for voltage division, connected between the internal reference voltage E6 and the signal input terminal for current detection IS, as well as Zener diodes D5, D6 which ground the connection point between the resistors R13 and R14. Here, the negative-voltage current detection signal (the signal itself is a voltage) VIS applied to the signal input terminal IS outputs to the current comparator 17 as an internal signal VIS2, which has been level-shifted to a positive potential, from the connection point of the resistors R14 and R15.
In this way, the current detection signal VIS is supplied to the signal input terminal for current detection IS as a negative voltage (0 to −1 V); because the IC circuit, which does not have a negative-voltage supply, cannot actually handle a negative-voltage signal, the level shift circuit 24 of FIG. 15 shifts the signal level to a positive potential (2 to 1.5 V).
At this time, the resistance values of the resistors R11, R12 and the like are adjusted such that the output level conforms to this current detection signal, even for the signal inversion circuit 23 which processes the feedback signal VFB.
Next, the function of the correction resistor Rc is explained. The correction resistor Rc adds a positive (positive voltage) offset voltage (bias) to the current detection signal VIS, so that in effect the switching frequency determined by the integrated circuit IC is lowered, in order to reduce the power consumption during light loading or no loading or the standby power during standby. Below, the manner in which the correction resistor Rc lowers the switching frequency is explained.
FIGS. 16(A)-16(C) are signal waveform diagrams explaining the correction operation of the current detection signal VIS in a switching power supply device. Here, the signal VFB3 is a hypothetical signal used for explanation, and is equivalent to the above-described internal signal VFB2, with operating range in the positive voltage range (for example 2 to 1.5 V), level-shifted such that the upper limit is 0 V to conform with the operation range of the current detection signal VIS, which is a negative voltage. It may be regarded as the result of inversion of the feedback signal VFB.
Here it is assumed that the oscillation frequency of the voltage controlled oscillator 14 is controlled by the feedback signal VFB supplied to the power supply control circuit 10.
First, as shown in FIG. 16(A), cases in which correction by the correction resistor Rc is not performed are considered. At this time, the turn-on time ratio of the switching element Q1 and the value of the feedback signal VFB are in a state of balance such that the voltage output to the load Vout is at the voltage setting. The switching frequency is then determined by the magnitude of the feedback signal VFB.
Next, suppose that the correction resistor Rc is added and correction is suddenly applied to the state in FIG. 16(A). In this case, the current detection signal VIS is a signal which starts to decline from a larger positive voltage than that in FIG. 16(A). On the other hand, the feedback signal VFB, that is, the signal VFB3 in the figure, cannot change rapidly, so that the same voltage level continues for a time. The switching element Q1 is not turned off until the current detection signal VIS reaches VFB3, so that as shown in FIG. 16(B), the turn-on time ton of the switching element Q1 is lengthened (the turn-on time ton is the interval from the time at which the current detection signal VIS begins to decline until the signal VFB3 is reached). At this time, if the switching frequency remains unchanged, then the turn-off time within one period is shortened, and the turn-on ratio of the switching element Q1 is increased. As a result, the voltage output to the load rises, the feedback signal VFB is reduced, and the absolute value of the feedback signal VFB3 is also reduced.
When the feedback signal VFB becomes small, the switching frequency declines, and the time ratio falls, so that the initial turn-on time ratio shown in FIG. 16(A) is approached. Hence as shown in FIG. 16(C), there is balancing at a new switching frequency, and finally it becomes the same turn-on time ratio as in FIG. 16(A). At this time, the feedback signal VFB and the absolute value of the shifted voltage value VFB3 are smaller than the values before correction. In this way, the frequency controlled by the voltage controlled oscillator 14 goes lower, and the turn-on time determined by the current comparator 17 is also lengthened.
In the above-described negative-detection type switching power supply device, current flowing in the correction resistor Rc during light loading remains a problem with respect to promoting energy efficiency. That is, because the correction resistor Rc in a switching power supply device of the prior art is connected to the power supply terminal VCC, current always flows from the power supply terminal VCC through the correction resistor Rc and the series circuit of the resistors Ra and Rs, and through the correction resistor Rc and the series circuit of the resistor Rb and auxiliary windings Lb, to ground (GND), so that there is the problem of the occurrence of power losses.
Offsets from two sources are applied to the current detection signal VIS: one from the output voltage of the auxiliary windings Lb via the resistor Rb, and another from the voltage of the power supply terminal VCC via the correction resistor Rc. The voltage of the power supply terminal VCC is proportional to the output voltage Vout, and the output voltage Vout is controlled so as to be a constant voltage, so that the voltage of the power supply terminal VCC is also a constant voltage. On the other hand, because the output voltage of the auxiliary windings Lb is proportional to the input voltage VIN, the value essentially fluctuates. Hence there is the problem that the correction resistor Rc has a complex effect on overcurrent detection.
That is, overcurrent detection is performed by comparing the voltage signal from the current detection input terminal IS to a certain reference voltage; however, it is difficult to adjust the circuit constants such that the comparison provides a constant result, regardless of the value of the input voltage VIN. This is because the voltage of the auxiliary windings Lb, which is proportional to the input voltage VIN, is applied to one end of the resistor Rb, and this resistor Rb is connected to the input terminal IS in parallel with the correction resistor Rc, to one end of which is applied a constant voltage (the power supply voltage VCC of the power supply control circuit 10, a regulated voltage), so that both affect the current detection signal VIS, and constantly adjustment of circuit constants is difficult.
This invention has been made in light of the above problems, and has as an object of the provision of a switching power supply device in which, when an external correction circuit is added and the switching frequency during light loading is adjusted from outside, losses in the correction circuit are reduced as compared with the prior art, and moreover adjustment is possible without affecting overcurrent limits or other characteristics.
Further objects and advantages of the invention will be apparent from the following description of the invention.